1. Field of the Invention
The present invention relates to predistorting signals in order to compensate for non-linearities, and in particular to linearization for small signal receiving devices or high power transmitting devices, operating at radio frequencies.
2. Description of Related Art
Communications systems handling a single carrier, usually employ high efficiency and low linearity (class C) amplifiers. Multi-carrier systems, however, require highly linear (class A) amplifiers at the expense of efficiency. An ideal amplifier would exhibit both perfect linearity and high efficiency.
Intermodulation distortion (IMD) products are produced by non-linearities in amplifiers handling radio frequency signals, such as the multiple carrier signals found in cellular telephone systems or in various other types of personal communications systems (PCS). Distortion can be caused by amplitude compression or phase shifts that occur for relatively large amplitude signals. The resulting spurious signals are added to the spectrum of the information bearing signals and thus degrade the quality of associated communications.
While the phenomena are fairly complex, the transfer function of an amplifier exhibiting non-linearities can be approximated by a Taylor series, that is, a polynomial with terms of the form a.sub.n x.sup.n. The first order effect (ax) is the desired linear function. Distortion is caused by the second order term (square term), the third order term (cubic term), and so forth.
For many high frequency systems, the even order terms produce harmonics that are out of the working bandwidth of the system and therefore rejected. For example, in a two carrier system with carrier frequencies f.sub.1 and f.sub.2, the second order harmonic frequencies are 2f.sub.1 and 2f.sub.2. These harmonic terms are typically far removed from the spectrum of the two carriers and can easily be removed by filtering.
For third order effects, the harmonic components of distortion include frequencies at 2f.sub.1 -f.sub.2, and 2f.sub.2 -f.sub.1, which are typically near the spectrum of the main carrier components. Other third order components are far outside the working spectrum and can be easily filtered. Typically, only odd order (third, fifth, seventh, etc.) intermodulation products are a concern.
Conventional amplifiers when driven to only modest power output operation levels well defined regions where output power of the odd harmonics bear a simple relationship to the power input. For example, the third order components, measured as a ratio to power input in dB, will increase at a rate three times the rate of increase of input power. Thus, a one dB increase in input power will cause a three dB increase in the third order distortion component d.sub.3 (FIG. 5, also showing the overall transfer function G on a log-log scale of power input P.sub.i vs. power output P.sub.o). At the same time the fifth order component will increase by five dB, the seventh order component seven dB, etc.
This orderly relationship between the change in the power of distortion components with input power breaks down for relatively high or relatively low power levels where the distortion curve flattens. The changes in distortion in this poorly defined region depend on device characteristics and other subtle phenomena making compensation difficult.
A linearizer may be used to reduce the effects of intermodulation distortion products. Am improvement is measured as an increased carrier to intermodulation power ratio (C/I). The two most common uses for linearizer networks are to improve the C/I performance of either small signal receiving amplifiers or high power transmitting devices, both of which are found in satellite communications systems. For small signal devices, linearization improves the overall dynamic range of the receiving system. Thus, the cost of a linearizing network is overcome by the significant C/I performance improvement. When a linearized amplifier is compared to a device, without linearization but with an equivalent performance, the savings can become substantial. The overall energy consumed by the linearizing network in receiving systems is significant but is not of concern due to the small power levels.
High power class A amplifiers are inefficient and can be very expensive. Benefits of linearization are realized as an improved IMD performance with less energy and cost being expended to attain a performance equivalent to that of a higher power and, consequently, more expensive device. The additional power dissipation of the linearizing network is normally a small fraction of the overall amplifier power consumption.
Effective power can be used as a measure of the relative merit of linearizing a device. A highly efficient device exhibits low power consumption and high output power for a given linearity. The overall "effective" efficiency is obtained by comparing the power consumption of a linearized amplifier to that of a standard amplifier of equivalent performance. If only third order IMD products are considered significant, then a 9 dB C/I improvement would equal the performance of a device more powerful by a factor of 3 dB. The general relation between equivalent power performance and improvement in C/I provided by a linearizer is indicated by the equivalent power performance (EPP), which equals 1/3 [C/I] (in dB). The "effective" efficiency factor is defined as 10 exp(EPP/10).
The previous example (9 dB improvement in C/I) translates into an increased "effective" efficiency of 2 times the existing (non-linearized) efficiency. The non-linearized efficiency is defined as the ratio of output power to input power (P.sub.o /P.sub.i). Maximum theoretical efficiency of a class A amplifier occurs at maximum power output (compression) and is equal to 50%. See J. Millman, "Microelectronics: Digital and Analog Circuits and Systems," McGraw-Hill Inc., New York, N.Y., pp. 666-667 (1979).
Several linearization approaches are available. These include feedback, feed forward, and predistortion. The first two are considered more complex and, consequently, more expensive than the third--predistortion. For feedback linearization in general, see E. Ballesteros, F. Perez and J. Perez, "Analysis and Design of Microwave Linearized Amplifiers Using Active Feedback," IEEE Trans. on Microwave Theory Tech., Vol. 36, No. 3, pp. 499-504; March 1988.
The second technique, feedforward linearization, involves extracting the actual distortion produced by an amplifier or other device. The extraction is performed by using the unamplified signal to cancel the undistorted signal component produced by the amplifier, leaving only distortion. This distortion is then boosted and used to cancel distortion in the amplified signal. In U.S. Pat. No. 5,304,945 two such stages of feedforward, distortion compensation are employed. See also U.S. Pat. Nos. 4,879,519 and 4,885,551. For a general discussion of feedforward linearization, see M. Sidel, "A Microwave Feed-Forward Experiment," Bell System Technical Journal, Vol. 50, pp. 2861-2879; September 1971.
These feedforward techniques may also employ a locally injected pilot signal used to facilitate feed back control of the distortion cancellation process. See for example, U.S. Pat. Nos. 4,580,105; 5,130,663; and 5,155,448.
With the third technique (predistortion) signals applied to the input of an amplifier are predistorted in such a way that the intentionally added distortion effectively cancels the distortion generated by the amplifier itself. For example in U.S. Pat. No. 4,588,958 a predistortion circuit has a linear and non-linear terminator connected through a directional coupler to the input of an amplifier to be linearized. The non-linear terminator can include diodes for generating distortion. In U.S. Pat. No. 4,882,547 a predistortion circuit of that type is automatically adjusted based upon a measurement of the ratio of the carrier to second harmonic.
Predistortion circuits have used an FET linearizer for generating predistortion for compensating an amplifier. Devices of this type essentially employ a single, forward signal path with the FET generating compensating distortion prior to driving the amplifier. See for example U.S. Pat. Nos. 5,038,113; 5,162,748; 5,191,338; and 5,138,275. In U.S. Pat. No. 4,564,816 a pair of FETs are both used to produce distortion in a predistortion circuit. See also U.S. Pat. No.4,488,122 for generation of distortion using a ferrimagnetic material.
In U.S. Pat. No. 4,554,514, a predistortion circuit has an adjustable phase shifter and adjustable attenuator coupled to the input of a power amplifier. The output of that power amplifier is coupled in a digital feedback loop to adjust the phase and attenuation produced by the predistortion circuit.
In U.S. Pat. No. 4,772,855 a predistortion signal is produced by a circuit having adjustable phase and amplitude modulators. These two modulators are controlled by a circuit that senses the amplitude of the input to the modulators to feed forward a signal for controlling both modulators. This system effectively uses a feed forward circuit to produce a non-linear, predistortion signal.
U.S. Pat. No. 5,361,156 shows a predistortion circuit having a linear branch and a branch with a distortion generator. The branch with the distortion generator has adjustable amplitude and phase. The predistortion is produced to compensate for distortion occurring in an optical communications link. Another system having a linear and non-linear branch is shown in U.S. Pat. No. 5,304,944. Neither of these systems attempt to isolate the distortion for separate treatment. Instead, the distortion generator produces distortion that remains combined with the undistorted, carrier component. See also U.S. Pat. No. 5,396,190.
In U.S. Pat. No. 4,987,378 two signal samplers (dividers) cooperate with two signal combiners to establish two signal loops. One of the loops has a pair of branches: One branch has a variable phase adjustment and another branch has a distortion generator cascaded with circuits that adjust phase and amplitude. These branches produce through one of the signal combiners a signal that has only distortion, that is, the carrier signal is removed. This pure distortion signal is then recombined with an the original, undistorted signal using phase adjustment and amplitude adjustment to produce a predistortion signal.
A disadvantage with circuits of this type is the lack of tracking between different branches designed to produce the distortion signal without the carrier. For example, U.S. Pat. No. 4,987,378 mentions building the distortion generator branch either with anti-parallel diodes or an amplifier that is driven into compression. The other branch, used to cancel the carrier signal, has very different structure: an adjustable phase shifting circuit employing a hybrid coupler connected to varactor diodes to produce an adjustable phase shift. The difficulty with using a mixture of circuits of various types is the inability of the two branches to accurately track in phase and distortion as the amplitude of the drive signals through the two branches varies.
Moreover, because of the overall circuit topology, the linearizer circuit U.S. Pat. No. 4,987,378 has multiple phase adjustments. In fact, each disclosed embodiment is shown with three or four independent adjustments. This large number of adjustments clearly complicates the use of the linearizer in practical embodiments.
For other references concerning predistortion and other techniques, see:
A. M. Killia, "Linearizers for Microwave Power Amplifiers in Communications Satellites," ANT Nachrichtentechnik (1988).
A. Katz, R. Sudarsanam and D. Aubert, "A Reflective Diode Linearizer for Spacecraft Applications," IEEE Trans. on Microwave Theory and Tech., pp. 661-664 (June 1985).
M. Kumar, J. C. Wartenby and H. J. Wolkstein, "Predistortion Linearizer Using GaAs Dual-Gate MOSFET for TWTA and SSPA Used in Satellite Transponders," IEEE Trans. on Microwave Theory and Tech., Vol. MTT-13, No. 12, pp. 1479-1488 (December 1985).
D. Cahana, J. R. Potukuchi, R. G. Marshalek and D. K. Paul, "Linearized Transponder Technology for Satellite Communications Part 1: Linearized Circuit Development and Experimental Characterization," COMSAT Technical Review, Vol. 15, No. 2A. pp. 277-308 (Fall 1985).
D. Pham, G. Lindgren and J. Steck, "A C-Band TWT Linearizer for Satellite Up-Link Transmitters," IEEE MTT-S International Microwave Symposium (May 1988).
G. Satoh, "MIC Linearizer for Satellite Communications," IEEE MTT-S International Microwave Symposium (May 1988).
T. Nojima, "Linearizers for Microwave Power Amplifiers," IEEE MTT-S International Microwave Symposium (May 1988).
T. C. Edward, "Foundations for Microstrip Circuit Design," John Wiley and Sons, New York, N.Y., pp. 57-59 (1985).
E. C. Jordan, "Reference Data for Engineers," Howard W. Sams and Co., Inc., Indianapolis, Ind., pp. 6.2-6.4 (1985).
H. Howe, Jr., "Stripline Circuit Design," Artech House Inc., Dedham Mass., pp. 261-266 (1985).
G. L. Matthaei, L. Young and E. M. T. Jones, "Microwave Filters, Impedance-Matching Networks, and Coupling Structures," Artech House Inc., Dedham Mass., pp. 755-809 (1980).
H. Howe, Jr., "Stripline Circuit Design," Artech House Inc., Dedham Mass., pp. 111-180 (1985).
J. P. Shelton, Jr., "Impedances of Offset Parallel-Coupled Strip Transmission Lines," IEEE Trans. on Microwave Theory and Tech., Vol. MTT-14, No. 1, pp. 715 (January 1966).
V. F. Fusco, "Microwave Circuits, Analysis and Computer-Aided Design," Prentice-Hall International (UK) Ltd., pp. 316-322 (1987).
H. Howe, Jr., "Stripline Circuit Design," Artech House Inc., Dedham Mass., pp. 77-110 (1985).
An object of the present invention is to provide an improved linearizer combining feedforward techniques with predistortion.